Frequency hopping code division multiple access radio communication unit

ABSTRACT

A radio communication unit for a digital communication system is provided in which an input information signal is protected from transmission errors by forward error correction encoding the information signal. In addition, the communication unit enhances subsequent processing of a transmitted form of the information signal by a hard-limiting receiver by inserting a predetermined synchronization sequence into the information signal. Further, a corresponding radio communication unit is provided which includes a hard limiting mechanism for removing the magnitude of each sample in a group of data samples of a signal received from over a radio communication channel. In addition, weighting coefficients of the hard-limited group of data samples for maximum likelihood decoding and diversity combining are generated by comparing the hard-limited group of data samples to a known predetermined synchronization sequence. Finally, estimated information samples are generated, utilizing the weighting coefficients, by maximum-likelihood decoding the group of data samples.

RELATED INVENTIONS

The present invention is related to the following inventions which areassigned to the assignee of the present invention:

Dual Mode Communication Network by Morton Stern et al. having U.S. Ser.No. 07/906,785, and filed on Jun. 30, 1992.

Method Of Registering/Reassigning A Call In A Dual Mode CommunicationNetwork by Borth et al. having U.S. Ser. No. 07/957,122, and filed onOct. 7, 1992.

FIELD OF THE INVENTION

The present invention relates to radio communication systems and, moreparticularly, to a frequency hopping code division multiple access radiocommunication unit.

BACKGROUND OF THE INVENTION

Cellular radio communication systems typically include a number ofcentral communication base sites. Each central communication site has aservice area coverage for servicing mobile communication units withinthe service area. The service areas typically are arranged such thatadjacent remote base site service coverage areas overlap in a mannerthat provides a substantially continuous service region. Thesubstantially continuous service region provides uninterrupted serviceby handing off mobile communication units from one base site serving aservice area to an adjacent base site serving another service area.

Pedestrian as well as mobile users will typically access the samecellular radio communication systems. For purposes of this discussion, apedestrian user is one who roams slowly (10 kph, kilometers per hour, orless) as opposed to a mobile user (up to 100 kph or more) user. However,these cellular communication systems are typically designed to provideadequate performance for the worst case environment (i.e., the mobileuser). As such, the cellular radio communication systems typicallyprovide continual overhead measurements used by the system to maintainchannel quality or perform hand-off functions. Since these measurementsrequire the same amount of processing whether a user is a mobile user ora pedestrian user, the pedestrian user is charged the same fee for usingtheir cellular phone as the user who is a mobile user.

Therefore, them exists a need in the industry for a personalcommunication system (PCS) which would provide a low-tier system forpedestrian users at a reduced cost. The low-tier system would provideaccess via radio frequency (RF) link to a basic cellular network whichmay or may not provide hand-off capability between low-tier serviceareas. In addition, a high-tier system should be provided for the mobileuser. This high-tier system would have many of the features found incurrent cellular systems including hand-off between high-tier serviceareas.

It is desirable to provide a high-tier PCS communication unit designwhich capable of performing all of these features by expanding uponlow-tier PCS communication unit designs. This high-tier PCScommunication unit design attempts to minimize cost, power consumption,and complexity, while maximizing RF spectrum usage per channel androbust design features (e.g., compact and integrated design) for highvolume manufacturing of the communication units.

SUMMARY OF THE INVENTION

A radio communication unit for a digital communication system isprovided in which an input information signal is protected fromtransmission errors by forward error correction encoding the informationsignal. In addition, the communication unit enhances subsequentprocessing of a transmitted form of the information signal by ahard-limiting receiver by inserting a predetermined synchronizationsequence into the information signal. Further, a corresponding radiocommunication unit is provided which includes a hard limiting mechanismfor removing the magnitude of each sample in a group of data samples ofa signal received from over a radio communication channel. In addition,weighting coefficients of the hard-limited group of data samples formaximum likelihood decoding and diversity combining are generated bycomparing the hard-limited group of data samples to a knownpredetermined synchronization sequence. Finally, estimated informationsamples are generated, utilizing the weighting coefficients, bymaximum-likelihood decoding the group of data samples.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram showing a preferred embodiment frequency hoppingcode division radio communication unit.

FIG. 2 is a diagram showing an alternative preferred embodimentfrequency hopping cede division radio communication unit.

DETAILED DESCRIPTION

Referring now to FIG. 1, a preferred embodiment high-tier PCScommunication unit 100 is depicted in block diagram form. As shown, thecommunication unit 100 may be logically separated into transmitter 102and receiver 104 function portions. It will be appreciated by thoseskilled in the art that although these communication functions have beenlogically separated, the actual implementation of these functions may beaccomplished in a variety of different manners including, but notlimited to properly programming a digital signal processor (DSP),coupling discrete components together, and using a combination of one ormore application specific integrated chips (ASICs). The transmitterportion 102 receives an information signal 106. The information signal106 may contain data or digitized speech. In the case that theinformation signal 106 contains digitized speech, the information signal106 is processed by a speech coder 108 to further encode the digitizedspeech. Preferably this speech coder employs a voice activity detection(VAD) mechanism to minimize the number of encoded data bits 110 whichrepresent the digitized speech. In the alternative, if the informationsignal 106 contains data, then the data is passed through the speechcoder 108 as the encoded data bits 110.

These encoded data bits are subsequently coded 112 with error detectionand error correction codes. In the preferred transmitter portion 102design, a cyclic redundancy code (CRC) is used for error detection, anda convolutional code is used for forward error correction. The length ofthe CRC is chosen such that it will reliably detect errors while notbecoming computationally burdensome. A similar strategy is followed inchoosing the constraint length of the convolutional codes. Thecomplexity of the code may be tailored to the particular hardwareimplementation. In addition, the high-tier PCS transceiver 100 mayoptionally utilize non uniform coding rates and selective application ofCRC in error detection coding.

After coding, the data bit stream is interleaved 112 to dispersetransmission errors of a single frequency hop over a larger period. Thehigh-tier communication unit 100 preferably implements a convolutionalinterleaver, because this structure results in a dispersion of channelerrors which is superior to a block interleaver having twice the amountof interleaving delay. By choosing the vertical dimension of theinterleaver to be evenly divisible into the number of interleavedsymbols transmitted within a frequency hopping slot, the interleaver hasbeen structured in a manner which will allow synchronization even if aslot is dropped. A known synchronization preamble is added 112 to theinterleaver output, and the data bit stream 114 is formatted for timedivision multiple access (TDMA) transmission. The data bit stream 114 isfiltered by a full raised cosine filter 116 with a rolloff factor of 0.5to meet bandwidth and intersymbol interference requirements.

This filtered data bit stream 118 preferably is subsequently four-phasemodulated 120 (i.e., quadrature phase shift keying (QPSK) modulated).The four-phase modulated data bit stream 122 preferably is provided toone input of mixer 124 and a frequency hopping carrier signal 126 isprovided to the other input of mixer 124. The frequency hopping carriersignal 126 preferably is generated by a frequency hop synthesizer 128which generates a carrier signal within a predetermined RF band thathops according to a predetermined pattern (i.e., the frequency hopsynthesizer 128 steps through the frequency hopping code). The mixedfrequency hopping signal 130 is subsequently amplified by poweramplifier 132, supplied 134 to a final stage filter 138 and radiated byantenna 138 over a communication channel.

It will be appreciated by those skilled in the art that thepredetermined RF band does not have to be a contiguous frequency band,but rather only need be within a specific range of frequencies to whichthe chosen frequency synthesizer is capable of operating. In addition,the predetermined pattern (i.e., hopping code) is used to determine thesequence in which a particular communication unit is to hop over the RFband such that the communication unit causes minimal interference toother communication units operating in the same multiple accesscommunication system. Further, it will be appreciated that frequency hoptransmission is employed to help mitigate channel impediments such asslow fading. Furthermore, the use of frequency hopping provides anotherform of diversity to the high-tier communication system and results inthe system performance being independent of a users speed (e.g., if auser is traveling in a vehicle).

The receiver portion 104 is designed to efficiently detect and decodethe transmitted signal. In view of the fading and multipath channelswhich are prevalent in mobile communications, a diversity receiver isemployed to improve performance. At the receiver 104, each diversitybranch (i.e. first branch 138, 142, 146, 150, 154, 158, 162, and 166 aswell as second branch 140, 144, 148, 152, 156, 160, 164, and 168) firstfilters 142, 144 and down converts 150, 152 its respective receivedsignal to a low IF frequency of approximately four megahertz. Thefrequency hop synthesizer 128 is used within the down conversion process150, 152 to follow the hopping signal. At this point the signal 154, 156is hard limited 158, 160. This feature eliminates the need for any formof automatic gain control (AGC) and greatly reduces the requiredresolution of the analog to digital (A/D) converter and the size of thedata paths required in the digital portion of the receiver 104. Thealgorithms and techniques which allow the use of a hard limiter 158, 160in a coded system are some of the most innovative and valuable featuresof the high-tier PCS communication unit 100.

Following the RF and IF processing, the low IF signal is bandpasssampled and converted to the digital domain 158, 160. A relatively lowcost A/D converter preferably samples at sixteen times the symbol rateand has only four bits of resolution. Four bit quantization can be used,because the earlier hard limiting 158, 160 has removed the magnitude ofthe four-phase waveform. At this point, additional filtering isperformed by low complexity (e.g., three to five taps) digital bandpassfilters to eliminate DC offsets, reduce sampling noise, and separate theinphase and quadrature branches. Each branch may then be decimated 158,160 by a factor of four effecting a translation to baseband of thehard-limited data samples. The translation to baseband can be readilyperformed, because the careful selection of a low IF frequency allowsthe communication unit 100 to utilize the image frequencies.

Now each branch is correlated 166, 168 with the known predeterminedsynchronization word to determine the optimum sampling point and toperform carrier recovery. Preferably the transmitted signal structurehas the synchronization word inserted before the data such that thecorrelation 166, 168 can be performed with only minimal buffering of thereceived signal 162, 164. The largest correlation magnitude can serve asan estimate of the channel gain, and the phase of this correlationreflects the conjugate of the phase correction required by the signal162, 164. Once the largest correlation has been determined, the datasamples are further decimated to single sample per symbol. Preferably,the high-tier communication unit 100 actually performs very littleprocessing with oversampled digital data. This allows the receiverportion 104 to minimize power consumption, memory storage and cost.

Next, a signal quality estimate, or weighting parameter 176, 178, iscalculated 174 for each branch 170, 172, and the branches 180, 182 arediversity combined 164. Within this process, the scaling 176, 178required for soft derision decoding 202 is also applied 164 to thesignal 180, 182. While it is possible to use a variety of weightingparameters, the best performance will be obtained from a ratio-basedstatistic. The ability of the high-tier communication unit 100 tocompute a ratio-based statistic allows the successful calculation ofsoft information despite the presence of the hard limiter 158, 160.

Since the ability to calculate 174 accurate weighting parameters 176,178 via low complexity techniques is crucial to the operation of thehigh-tier PCS, the derivation of the weighting parameters 176, 178 shallbe discussed in detail. In order to establish a framework for thisdiscussion, it is assumed that an arbitrary binary communication channelwith time-varying channel gain and noise variance can be modeled as

    r=p.sub.o x.sub.s +n                                       (eq. 1)

where r is the received signal vector, p_(o) is the channeled gainmatrix, x_(s) is the transmitted signal vector, and n is the noisevector. Each element of x_(s), denoted as x_(s) (k), is an independentidentically distributed binary random value taking values ±√c with equalprobability, and each element of n is an independent Gaussian randomvariable with zero mean and variance σ_(n) ² (k). Thus, the optimumsignal weighting for the maximum likelihood decoder 202 may be writtenas ##EQU1##

Furthermore, this weighting coefficient 176, 178 may serve as theoptimal max ratio diversity combining coefficient. Thus, within thisframework, the computation of the soft derision weighting and diversitycombining coefficients 176, 178 reduces to the calculation of a singlecoefficient formed from the ratio of the channel gain to the noisevariance. For the preferred embodiment high-tier PCS, this basic modelis valid with the qualification that the received signal power isapproximately constant due to the effects of the hard limiter 158, 160.Since the limiter 158, 160 has an equal effect on the desired signal andthe noise, the ratio derived above still serves as a valid estimate ofthe signal reliability.

At the receiver portion 104, only the received signal r(k) 170, 172 isavailable. Noting that r(k) has zero mean, it is possible to define thereceived signal variance as ##EQU2## Taking the expectations, recallingthat n(k) and x_(s) (k) are independent and zero mean, yields

    σ.sub.r.sup.2 (k)=cp.sub.0.sup.2 (k)+σ.sub.n.sup.2 (k). (eq. 4)

If the error signal is defined as e_(s) (k)=r(k)-x_(s) (k), then in thesame manner as for the received signal variance, the variance of theerror signal may be determined as ##EQU3## Now a straightforwardalgebraic manipulation of (eq. 4) and (eq. 5) yields ##EQU4## and usingthis result

    σ.sub.n.sup.2 (k)=σ.sub.r.sup.2 (k)-cp.sub.0.sup.2 (k). (eq. 7)

While (eq. 6) and (eq. 7) provide a means for calculating the componentscomprising the weighting parameter, by considering the calculation ofthe error variance in slightly different manner, a lower complexitymethod which is ideally suited to the high-tier PCS can be obtained. Inthis case, the error variance is expanded as ##EQU5## where R_(rx)(k,k), hereafter denoted by R_(rx), represents the cross correlationbetween the received and transmitted signals. In general, this crosscorrelation would be of little use since the transmitted signal is notavailable at the receiver. In high-tier PCS, however, a ten symboltraining sequence is preferably incorporated into the beginning of eachtransmission slot. Since these symbols are known and the length of thesequence is sufficiently long, the correlation may be accuratelycalculated.

Substituting (eq. 8) into (eq. 6) and simplifying yields ##EQU6##

This new solution for the channel gain may then be substituted into (eq.4) to yield

    σ.sub.r.sup.2 (k)=c(R.sub.rx /c).sup.2 +σ.sub.n.sup.2 (k). (eq. 10)

Solving (eq. 10) for σ_(n) ² (k) and using (eq. 9), allows the solutionof (eq. 2) as ##EQU7##

This solution for the diversity combining and soft decision scalingcoefficient 176, 178 is well suited for implementation in the high-tierPCS. The cross correlation will routinely be calculated in hardware(e.g., in a prototype system this quantity was calculated by a properlyprogrammed field programmable gate array) as part of the timing andcarrier recovery process, and the received signal variance, which issimply the received signal power, is also readily obtainable. While thelimiter 158, 160 will normalize the received power to a constant, thedigital filters 158, 160 following the analog to digital conversionprocess will introduce a small data-dependent fluctuation into thevariance of the received signal. For this reason, σ_(r) ² (k) ismaintained in the denominator of the optimal solution indicated in (eq.11). In practice, however, this fluctuation is relatively small, andcσ_(r) ² (k) can be replaced by a fixed constant, α, with a negligibleloss in performance. In this case, the weighting parameter 176, 178would be represented by ##EQU8## and is a function only of the crosscorrelation.

It will be appreciated by those skilled in the art that a correlationfunction has been described above for calculating the weightingcoefficients; however, other types of comparison operations could beused such as mean squared error functions to perform this comparisonwithout departing from the scope and spirit of the present invention.

As FIG. 1 indicates, an equalizer 198 is a possible receiver portion 104option. Such an equalizer 198 would require input 194, 196 from eachbranch as well as the output 186 of diversity combiner 184. In addition,in order for the equalizer 198 to perform optimally, the equalizer 198would need to output fine tuning adjustment information to each branch190, 192 as well as the diversity combiner 188. Subsequently, theequalized combined data sample stream would be output 200 to thedeinterleaver 202. While high-tier PCS does not exclude the use of anequalizer 198, preliminary results indicate that its use results in onlya small performance improvement relative to a system employing onlyfrequency-hopping and diversity.

As an alterative to an equalizer 198, the synchronization word may beused to sound the channel thereby allowing the implementation of amatched filter receiver (shown in FIG. 2). On multipath channels, theuse of a matched filter 181, 183 may allow the recovery of a significantportion of the energy in the secondary rays without the full complexityrequired by an equalizer 198. Also, although not indicated in the blockdiagram, frequency control information will be developed from thediversity combined signal 186 and used to control the frequency hopsynthesizer 128 and down conversion process 150, 152.

Following the weighting and combining process 184, the data samplestream 186, 200 is convolutionally deinterleaved 202. The deinterleaveroutput is Viterbi decoded 202 in an attempt to correct the errorsintroduced by the communication channel. At this point, the errordetection code (e.g., cyclic redundancy check (CRC) code) may optionallybe used 202 to check for errors over the span of the code's input. Thedecoded bits 204, along with the CRC-derived erasure information,preferably is then output as data 208 or input to the speech decoder 206and then output as voice 208.

Alternatively, the preferred embodiment communication unit 100 shown inFIG. 1 can be described as follows. A radio communication unit 100 for adigital communication system having a transmitter portion 102 isprovided. The transmitter portion 102 includes data bit coder 108 forencoding a received information signal 106 into a data bit stream 110.The data bit coder 108 preferably encodes the received informationsignal 106 into a data bit stream 110 according to a information signalcoding algorithm and provides the data bit stream 110 to the errorcontrol mechanism 112 for subsequent forward error correction encoding.The information signal coding algorithm consists of:

(1) encoding the information signal 106 with a speech coding algorithmhaving voice activity detection, when the information signal 106includes digitized voice signals, and

(2) passing the information signal 106 through the data bit coder 108without additional coding, when the information signal 106 includes datasignals.

The data bit stream 110 is input to an error control mechanism 112 whichprotects the data bit stream 110 from transmission errors byconvolutionally encoding and interleaving the data bit stream. Thisprotected data bit stream 110 subsequently has a predeterminedsynchronization sequence inserted into the error protected data bitstream such that subsequent maximum ratio diversity combining andmaximum-likelihood decoding of the transmitted signal by a hard-limitingreceiver 104 is enhanced.

The error protected data bit stream 114 is subsequently multi-phasemodulated 120 to generate a multi-phase (e.g., four phase) intermediatesignal 122 through the use of the error protected data bit stream 118.This the intermediate signal 122 is frequency translated 124 togenerating a radio frequency transmission signal 130 by combining theintermediate signal 122 with a radio communication channel selectingsignal 126 generated by a frequency hop synthesizer 128. Subsequently,an antenna 138 transmits the radio frequency transmission signal 130(i.e., after it is amplified 132 and filtered 136) over a radiocommunication channel.

A radio communication unit 100 for a digital communication system havinga receiver portion 104 is provided. A first 138 and second 140 antennareceives a signal from over a radio communication channel. This receivedsignal 146 is demodulated to generating a first 154 and a second 156group of data samples of the received signal at an intermediatefrequency corresponding to the signal received from the first 138 andthe second 140 antenna, respectively, through the use of a radiocommunication channel selecting signal generated by a frequency hopsynthesizer 128. These first 154 and second 156 group of data samplesare hard limited 158, 160 to remove the magnitude of each sample in thefirst 154 and the second 156 group of data samples. A subset of thehard-limited data samples of the first 154 and the second 156 group arefrequency translated to baseband frequencies 162, 164 by decimating thefirst 154 and second 156 group of samples in the time domain.

These subsets of the hard-limited data samples of the first 162 and thesecond 164 group are correlated to a known predetermined synchronizationsequence to independently determine an optimal sampling point for thefirst 162 and second 164 group of data samples to generate symbol ratedata samples of the first and second group and to determine channelsounding information.

From the hard-limited symbol rate data samples of the first 170 and thesecond 172 group weighting coefficient are generated 174 for diversitycombining and maximum likelihood decoding. The weighting coefficients176, 178 (λ) are preferably generated as a function of the followingalgorithm: ##EQU9## where, p₀ (k)=the channel gain estimate,

σ_(n) ² (k)=the channel noise variance,

R_(rx) =the cross-correlation between the received data symbols and theknown predetermined synchronization sequence wherein thecross-correlation is a pan of the determined channel soundinginformation,

σ_(r) ² (k)=the received signal variance, and

c=the expectation of the square of the transmitted data bit.

Alternatively, the weighting coefficients 176, 178 may be calculatedaccording to a substantially similar algorithm: ##EQU10## where, α=aconstant which approximates the hard-limited received signal variance.

These weighting coefficients 176, 178 are used to scale the first 180and the second 182 group of symbol rate data samples and to maximumratio combine the first 180 and the second 182 scaled symbol rate datasamples into a stream of combined data samples 186. The stream ofcombined data samples 186 is deinterleaved and maximum-likelihooddecoded 202 into estimated information samples 204.

In summary, the high-tier PCS communication unit 100 offers severalimprovements over known technology. The techniques developed to allowthe use of soft derision decoding and weighted diversity combining on ahard-limited signal offer large performance gains relative toconventional hard decision decoding and selection diversity techniques.Furthermore, the combination of these techniques with the use of a hardlimiter and low resolution analog-to-digital converter yieldsperformance which would have previously been unobtainable in a receiverof similar complexity.

Although the invention has been described and illustrated with a certaindegree of particularity, it is understood that the present disclosure ofembodiments has been made by way of example only and that numerouschanges in the arrangement and combination of parts as well as steps maybe resorted to by those skilled in the art without departing from thespirit and scope of the invention as claimed.

What is claimed is:
 1. A receiver of a radio communication unit for adigital communication system, comprising:(a) hard limiting means forremoving the magnitude of each sample in a group of data samples of asignal received from over a radio communication channel; (b) weightingcoefficient generation means, coupled to the hard limiting means, forgenerating weighting coefficients of the hard-limited group of datasamples for maximum likelihood decoding by comparing the hard-limitedgroup of data samples to a known predetermined synchronization sequence;and (c) error control means, coupled to the hard limiting means and theweighting coefficient means, for maximum-likelihood decoding the groupof data samples into estimated information samples by utilizing theweighting coefficients.
 2. The receiver of claim 1 further comprisingdemodulating means, coupled to the hard limiting means, for generatingthe group of data samples of the received signal at an intermediatefrequency corresponding to the signal received on an antenna through theuse of a radio communication channel selecting a signal generated by afrequency hop synthesizer.
 3. The receiver of claim 1 wherein weightingcoefficient generation means comprises(a) means for correlating thehard-limited group of data samples to the known predeterminedsynchronization sequence to determine channel sounding information; and(b) means for generating weighting coefficients (I) of the hard-limitedgroup of data samples as a function of the following algorithm:##EQU11## where, p_(o) (k)=a channel gain estimate, σ_(n) ² (k)=achannel noise variance, R_(rx) =a cross-correlation between the receiveddata symbols and the known predetermined synchronization sequencewherein the cross-correlation is a part of the determined channelsounding information, σ_(r) ² (k)=a received signal variance, and c=anexpectation of the square of the transmitted data bit.
 4. The receiverof claim 3 wherein the weighting coefficient generation means furthercomprises means for generating the weighting coefficients (I) as afunction of the following algorithm: ##EQU12## where, p_(o) (k)=achannel gain estimate,σ_(n) ² (k)=a channel noise variance, R_(rx) =across-correlation between the received data symbols and the knownpredetermined synchronization sequence wherein the cross-correlation isa part of the determined channel sounding information, a=a constantwhich approximates the hard-limited received signal variance.
 5. Thereceiver of claim 1 further comprising a speech decoding means, coupledto the error control means, for converting the estimated informationsamples into an analog speech signal.
 6. A receiver of a radiocommunication unit for a digital communication system, comprising:(a)hard limiting means for removing the magnitude of each sample in a firstand a second group of data samples of a first and a second signalreceived from over a radio communication channel, respectively; (b)weighting coefficient generation means, coupled to the hard limitingmeans, for generating weighting coefficients of the hard-limited datasamples of the first and the second group for maximum likelihooddecoding by comparing the hard-limited data samples of the first and thesecond group to a known predetermined synchronization sequence; and (c)diversity combining means, coupled to the hard limiting means and theweighting coefficient means, for scaling the first and second group ofdata samples and diversity combining the first and the second scaleddata samples into a stream of combined data samples.
 7. The receiver ofclaim 6 further comprising demodulating means, coupled to the hardlimiting means, for generating the first and the second group of datasamples at an intermediate frequency corresponding to the first andsecond signal received, respectively, from a first and a second antenna,respectively, through the use of a radio communication channel selectingsignal generated by a frequency hop synthesizer.
 8. The receiver ofclaim 6 wherein the weighting coefficient generation means comprises(a)means for correlating the hard-limited data samples of the first and thesecond group to the known predetermined synchronization sequence todetermine channel sounding information; and (b) means for generatingweighting coefficients of the hard-limited data samples of the first andthe second group as a function of the following algorithm: ##EQU13##where, p_(o) (k)=a channel gain estimate, σ_(n) ² (k)=a channel noisevariance, R_(rx) =a cross-correlation between the received data symbolsand the known predetermined synchronization sequence wherein thecross-correlation is a part of the determined channel soundinginformation, σ_(r) ² (k)=a received signal variance, and c=anexpectation of the square of the transmitted data bit.
 9. The receiverof claim 8 wherein the weighting coefficient generation means furthercomprises means for generating the weighting coefficients as a functionof the following algorithm: ##EQU14## where, P_(o) (k)=the channel gainestimate,σ_(n) ² (k)=the channel noise variance, R_(rx) =thecross-correlation between the received data symbols and the knownpredetermined synchronization sequence wherein the cross-correlation isa part of the determined channel sounding information, a=a constantwhich approximates the hard-limited received signal variance.
 10. Thereceiver of claim 6 wherein the diversity combining means comprisesmeans for diversity combining the first and the second scaled datasamples into a stream of combined data samples by maximum ratiocombining the first and the second scaled data samples into a stream ofcombined data samples.
 11. A receiver of a radio communication unit fora digital communication system, comprising:(a) hard limiting means forremoving the magnitude of each sample in a first and a second group ofdata samples of a first and a second signal received from over a radiocommunication channel, respectively; and (b) correlation means, coupledto the hard-limited means, for correlating the hard-limited data samplesof the first and the second group to a known predeterminedsynchronization sequence to determine channel sounding information; and(c) matched filter means, coupled to the correlation means, forincorporating energy of secondary rays of a multipath channel into thefirst and the second group of data samples by utilizing the channelsounding information to set filter coefficients of the matched filtermeans.
 12. The receiver of claim 11 further comprising demodulatingmeans, coupled to the hard limiting means, for generating for generatingthe first and the second group of data samples at an intermediatefrequency corresponding to the first and second signal received,respectively, from the first and the second antenna, respectively,through the use of a radio communication channel selecting signalgenerated by a frequency hop synthesizer.
 13. A radio communication unitfor a digital communication system, comprising:(a) antenna means,comprising a first and a second antenna, for receiving a signal fromover a radio communication channel; (b) demodulating means, coupled tothe antenna means, for generating a first and a second group of datasamples of the received signal at an intermediate frequencycorresponding to the signal received by the first and the secondantenna, respectively, through the use of a radio communication channelselecting signal generated by a frequency hop synthesizer; (c) hardlimiting means, coupled to the demodulating means, for removing themagnitude of each sample in the first and the second group of datasamples; (d) frequency translation means, coupled to the limiting means,for translating a subset of the hard-limited data samples of the firstand the second group to baseband frequencies by decimating the first andsecond group of samples in the time domain; (e) correlation means,coupled to the frequency translation means, for correlating the subsetof hard-limited data samples of the first and the second group to aknown predetermined synchronization sequence to independently determinean optimal sampling point for the first and second group of data samplesto generate symbol rate data samples of the first and second group andto determine channel sounding information; (f) weighting coefficientgeneration means, coupled to the correlation means, for generatingweighting coefficients of the symbol rate data samples of the first andthe second group, the weighting coefficients being generated as afunction of the following algorithm: ##EQU15## where, p_(o) (k)=achannel gain estimate, σ_(n) ² (k)=a channel noise variance, R_(rx) =across-correlation between the received data symbols and the knownpredetermined synchronization sequence wherein the cross-correlation isa part of the determined channel sounding information, σ_(r) ² (k)=areceived signal variance, and c=an expectation of the square of thetransmitted data bit; (g) diversity combining means, coupled to thecorrelation means and the weighting coefficient generation means, forscaling the symbol rate data samples of the first and the second groupand maximum ratio combining the first and the second scaled symbol ratedata samples into a stream of combined data samples; and (h) errorcontrol means, coupled to the diversity combining means, fordeinterleaving and maximum-likelihood decoding the stream of combineddata samples into estimated information samples.
 14. The radiocommunication unit of claim 13 wherein the weighting coefficientgeneration means generates the weighting coefficients (λ) as a functionof the following algorithm: ##EQU16## where, p_(o) (k)=the channel gainestimate,σ_(n) ² (k)=the channel noise variance, R_(rx) =thecross-correlation between the received data symbols and the knownpredetermined synchronization sequence wherein the cross-correlation isa part of the determined channel sounding information, α=a constantwhich approximates the hard-limited received signal variance.
 15. Theradio communication unit of claim 13 further comprising an equalizermeans, coupled to the correlation means, combining means, and errorcontrol means, for equalizing the first and second group of data samplesby incorporating energy of secondary rays of a multipath channel intothe first and the second group of data samples prior to the maximumratio combination of the first and the second group of symbol rate datasamples, and for outputting a combined- equalized stream of data samplesto the error control means.
 16. The radio communication unit of claim 13further comprising a speech decoding means, coupled to the error controlmeans, for converting the estimated information samples into an analogspeech signal.
 17. The radio communication unit of claim 13 furthercomprising a matched filter means, coupled to the correlation means andthe combining means, for incorporating energy of secondary rays of amultipath channel into the first and the second group of data samples byutilizing the channel sounding information to set filter coefficients ofthe matched filter means, and for outputting first and second group ofmatched-filtered data samples to the combining means.